Overcurrent limiter circuit for switching regulator power supplies

ABSTRACT

A circuit is disclosed which provides overcurrent protection for switching regulator power supplies with soft start and soft turn OFF features. The invention includes first means for sensing the current through the switching transistor of the regulator and second means for providing a voltage signal to the switching transistor deactivation circuitry whenever the current through the switching transistor exceeds a predetermined threshold. In a specific embodiment, the current generator is provided by a transistor biased for nominal operation in its active mode. The means for sensing the current through the switching transistor is provided by a resistor the voltage drop across which provides the input voltage threshold to the bipolar transistor current generator. The current generated by the transistor causes a voltage to quickly develop across an RC network referenced to ground potential. This voltage is then compared to a reference potential by a comparator circuit which provides an electrical signal as an input to the base drive circuit for the switching regulator switching transistor.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to switching regulator power supplies. Morespecifically, this invention relates to systems which provideovercurrent protection for such power supplies.

While the present invention is described herein with reference to aparticular embodiment in a particular application it is to be understoodthat the invention is not limited thereto. Those having ordinary skillin the art to which this invention pertains will recognize modificationsand other applications within the scope thereof.

2. Description of the Prior Art

As discussed by Abraham I. Pressman in Switching and Linear Power SupplyDesign, published by the Hayden Book Company in 1977, pages 183 through185, the routine testing and maintenance of power supplies is such thatthe output of power supply may be accidentally shorted to ground.Overload currents may occur from a breakdown or malfunction in the loadcircuit. While most overload currents are usually temporary, it isdesirable that they do not damage the power supply. For this reason,overcurrent protection circuits have been employed with some measure ofsuccess.

Prior art overcurrent protection circuits are primarily intended toprotect against over dissipating linear mode series pass elements whenan output node is shorted to ground. Although the prime reason for priorart overcurrent protection is to protect against such dead shorts at theoutput of the power supply, it is also useful to protect against smallerlimited overcurrents above the maximum specified load current.

Pressman discusses two generally used modes of overcurrent protection:constant current and current foldback. In a constant current protectionsystem, after output current is increased up to the maximum specifiedoverload current, the output current is held constant. In a currentfoldback system, the output voltage remains constant and withinspecification limits up to a predetermined current level. Beyond thiscurrent level, the output voltage and the current start falling backalong a "foldback" line.

For linear series pass regulators the current foldback technique mayhave advantages over the constant current technique. However, neither isdirectly applicable to a switching regulator type power supply. Priorart overcurrent limiters for switching regulators are adaptations ofthose previously developed for linear regulators. As such the prior artswitching regulator overcurrent limiter circuits depend on the resettingof a switching inductor core. This may be costly and otherwise difficultto achieve.

In addition, hysterisis type switching regulators (non-clocked) usingprior art overcurrent limiters, have a switching frequency that isrelated to the magnitude of the overload resistance. The switchingfrequency is very high for the "critical overload resistance" and lowestfor a short circuit. The "critical overload resistance" is defined hereas the load resistance slightly lower than the load resistance thatbarely trips the overcurrent limiter threshold setting.

The high switching frequencies obtained at the critical overloadresistance usually exceed the power switching frequency ratings of boththe transistor switch and the flyback diode. Thus, it is desirable toprovide a power supply having a switching frequency which is heldconstant and at the highest safe switching rate for all overloadresistances including the short circuit as well as the critical loadresistance.

In addition, it is desirable to provide a power supply with soft startand soft turn OFF performance characteristics. A soft start may berequired in the event that the load requires very high transientstarting current. Capacitors, motors, and incandescent lamps areexamples of loads requiring high transient starting current. A soft turnOFF may be required in the event that the source of unregulated power isswitched off which causes a switching regulator to sense a voltage dropwhich it tries to compensate for by drawing more current through theswitching transistor. In this mode, the unregulated power may bedecaying down due to the slow discharge of an input filter capacitor. Asa result, the power switch may have enough energy to be biased ON butnot enough to be in saturation. In that event very high current may bedrawn through the power switch in its active mode. This presents ahazard to a switching transistor without the soft turn OFF feature.

SUMMARY OF THE INVENTION

The present invention provides overcurrent protection for switchingregulator type power supplies with soft start and soft turn OFFfeatures. Generally, the invention includes first means for sensing thecurrent through the switching transistor of the switching regulator, andsecond means for providing an electrical signal to the switchingtransistor deactivation circuitry whenever the current through theswitching transistor exceeds a predetermined threshold.

In a specific embodiment overload current is sensed by a currentgenerator provided by a transistor biased to detect the overloadcurrent. The means for sensing the overload current through theswitching transistor is provided by a resistor, across which the voltagedrop provides an input to the overload current detector. The currentgenerated by the overload current transistor causes a voltage to developacross an RC integrator network which is referenced to ground level.This voltage is compared to a reference potential by an amplitudecomparator circuit which provides an electrical signal as an input tothe base drive circuit for the switching regulator switch transistor.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic diagram representation of a first embodiment ofthe present invention.

FIG. 2 is a schematic diagram representation of a second embodiment ofthe present invention.

FIG. 3 is a schematic diagram representation of a third embodiment ofthe present invention.

FIGS. 4a and 4b show voltage waveforms to further illustrate theoperation of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

The overcurrent protection circuit for switching regulator powersupplies of the present invention are shown in FIGS. 1-3.

FIG. 1 shows the first of three embodiments of the invention. Theovercurrent protection circuit of the present invention is showngenerally at 10 within dashed lines. A power switching transistor Q₁₁ isdriven by a base drive circuit 14. The switching transistor Q₁₁ isconnected between a source of unregulated line voltage V_(L) throughresistor R₁₂ and is connected to a load at V_(o) through inductor L₁₁. Aflyback diode CR₁₁ is connected to the junction between the switchingtransistor Q₁₁ and inductor L₁₁ and ground. A voltage sensing circuit 16is typically connected to V_(o) to monitor the output voltage. Theoutput of the voltage sensing circuit 16 is input to a hysteresisvoltage comparator 18 which drives the base drive circuit 14.

The base drive circuit 14, the voltage sensing circuit 16, and thecomparator 18 are shown in block diagram form to emphasize that thesecircuits may typically be provided in a switching regulator powersupply. The design of these circuits is known to one of ordinary skillin the art. The details of these block diagrams are not required for anunderstanding of the present invention. For illustrative designs seeU.S. Pat. Nos. 3,294,981 to Bose; 3,772,588 to Kelly et at; 3,931,567 toKosteck; 4,034,280 to Cronin et al; and Pressman, Supra, pp. 321-235.

As mentioned above, the overcurrent limiter circuit of the presentinvention is shown at 10. It includes a resistor R₁₂ in the current pathbetween the source of unregulated line voltage V_(L) and the emitter ofthe switching transistor Q₁₁. The invention 10 also includes anamplitude comparator transistor Q₁ connected so that it floats aboveground potential. The transistor Q₁ is base coupled to the emitter ofthe switching transistor Q₁₁ through a low pass filter, R₁ and C₁. Theresistor R₁₂ provides a current sensing resistor in that its voltagedrop is input to Q₁ through the R₁ /C₁ filter.

The resistor R₁ between the emitter of the transistor Q₁₁ and the baseof the transistor Q₁ and the capacitor C₁ across the base emitterjunction of the transistor Q₁ provides a low pass RC filter. This RCfilter removes from the base of the transistor Q₁ the large amplitude,fast (nanosecond) voltage spikes appearing across R₁₂ caused by the flowof current through CR₁₁ to ground at the instant Q₁₁ turns ON. ThoughCR₁₁ is typically a fast recovery diode, it has a finite recovery time.During its recovery time, it is in a conduction mode which permitscurrent to flow through the switching transistor Q₁₁ to ground. DuringCR₁₁ recovery interval, current surges through CR₁₁ as if it weremomentarily shorted. The low pass filter provided by the combination ofresistor R₁ and capacitor C₁ prevent these diode recovery current spikesfrom falsely triggering the overcurrent amplitude detector (currentgenerator) Q₁ . Using the component values shown in Table I below, theRC time constant of the filter is approximately 1 microsecond. As such,the filter does not appreciably delay inherent response time of theovercurrent limiter circuit 10.

Resistor R₂ is connected in the collector path of the transistor Q₁simply to provide parasitic oscillation suppression when Q₁ is in thelinear mode. In this configuration, Q₁ is both a floating overcurrentdetector and a current generator. Q₁ is either OFF when there is nooverload current or is in the linear mode when there is an overloadthrough the power supply switching transistor Q₁₁. Upon an overcurrentdetection, the Q₁ base voltage ramps linearly into the Q₁ active regionso that the Q₁ collector current resembles a linear current ramp. (To bemore precise, the collector current through Q₁ is nonlinear is Q₁ movesout of collector cut-off. However, for practical purposes, the Q₁collector current is a pseudo-linear increasing function.)

A capacitor C₂ is mounted in the collector path of transistor Q₁ toreceive current from Q₁ at ground potential. The resulting voltage dropacross the capacitor C₂ provides one of the differential inputs to acomparator circuit including amplifier U₁₁, resistors R₁₃ through R₁₈and capacitors C₁₁. Resistors R₁₅ and R₁₇ set the comparator thresholdlevel. Capacitor C₁₁ forms a low pass filter to minimize noiseinterference for the 2 level (hysteresis) reference potential ofcomparator U₁₁. R₁₄ establishes the amount of hysteresis of comparatorU₁₁ ; i.e., the upper and lower trip level (UT and LT) of comparatorU₁₁. R₁₃ is the pull-up resistor for the "wired OR" circuit at theoutput of comparator U₁₁ and block diagram 18.

The voltage drop across capacitor C₂ is applied to the comparator U₁₁through a voltage divider consisting of resistors R₁₆ and R₁₈. R₁₆ andR₁₈ also provide a discharge path for the capacitor C₂. As a result, thetime constant of the C₂ decay voltage is C₂ times the parallelcombination of resistors R₁₆ and R₁₈. C₂ charges quickly by integratingthe current ramp from Q₁ and discharges slowly through R₁₆ and R₁₈.

The capacitor C₂ integrates the linear collector current ramp from Q₁.The overcurrent comparator U₁₁ changes state after the voltage acrosscapacitor C₂ rises above its steady state value to the upper trip point(UT) of the comparator U₁₁. The change of the state of the comparatorU₁₁ initiates overcurrent turn OFF of Q₁₁ via the "wired OR" connectionto the base drive circuit 14. Upon turn OFF of Q₁₁, Q₁ turns OFF (inabout a microsecond) as soon as capacitor C₁ discharges a fraction of avolt. Q₁₁ and Q₁ remain OFF until the voltage across C₂ decays about onehalf volt (using the parameters specified in Table I below) which is thehysteresis of the comparator U₁₁. This charging and discharging of C₂and the comparator hysteresis form a relaxation oscillator to set theswitching regulator frequency and duty cycle during any overloadcondition.

The waveform occuring across C₂ for a continuous overload condition isshown in FIG. 4 for the parameters specified in Table I below. FIG. 4ashows the voltage waveform across capacitor C₂, and FIG. 4b shows thevoltage waveform at the comparator output. Note that the voltage dropacross the capacitor begins at the upper threshold (UT) of thecomparator U₁₁ (i.e., 3.35 volts) and decays of the U₁₁ lower threshold(UT) (i.e., 2.85 volts). It can be seen from FIG. 4b that during thetime the output of the comparator is low 0 volts, the base drive circuit14 is holding the switching transistor Q₁₁ OFF. However, when thevoltage drop across the capacitor C₂ reaches the lower threshold LT, theoutput of the comparator U₁₁ goes high, the base drive circuit goes lowand the switch Q₁₁ comes ON. Assuming that the overload or short circuitis still in place across V_(o) when Q₁₁ comes ON, current flowingthrough R₁₂ will again turn ON Q₁. When transistor Q₁ turns ON itcharges capacitor C₂ so that the output voltage of C₂ (see FIG. 4a)quickly reaches the new upper threshold at UT. This causes thecomparator U₁₁ to go OFF which causes the base drive circuit to go highthereby turning OFF the switching transistor Q₁₁.

The dotted line in FIG. 4a shows the decay path of the capacitor C₂. Thedistance h between the lower threshold LT and the upper threshold UT isthe hysteresis of the comparator of U₁₁. The pulse repetition frequencyfor the graph of FIG. 4b would be on the order of 1 kilohertz using thecomponents specified in Table I. The dash-dot vertical lines in FIG. 4show the time coincidences between FIGS. 4a and 4b.

The pulse width ΔT of FIG. 4b depends on the magnitude of the overloadresistance. For a short circuit, Δt will be minimum because the resetvoltage for inductor L₁₁ is minimum. The typical range for Δt is suchthat Δt is greater than 5 microseconds and less than 50 microseconds. ΔTis approximately equal to L₁₁ ×I₁ /(V₁ -V_(o)). Where L₁₁ is theinductance of the ##EQU1## the output voltage, and V₁ is equal to theunregulated input line voltage. It is seen from above that the PRF of1000 H_(z) is relatively independent of the magnitude of overloadresistance because 1/1000 H_(z) >>50 microseconds.

The frequency of the overcurrent comparator oscillation can easily beadjusted by changing the capacitance of C₂. The frequency should be setas high as possible and yet be within the maximum switching speed andfrequency ratings of Q₁₁ and CR₁₁. In the overcurrent limiting mode, theON duty cycle of the switching transistor Q₁₁ would be very low as shownin FIG. 4b. To stay within the finite switching speed limitations of Q₁₁and CR₁₁ the overcurrent limiting mode pulse repetition frequency shouldbe set low; i.e., about 1,000 H_(z).

Typical switching speed limitations for the transistor Q₁₁ and the diodeCR₁₁ require that the overcurrent recycle frequency be set about 1,000H_(z). Again, this recycle frequency is set via the capacitor C₂ timeconstant. Minimal overcurrent foldback improves the turn ONcharacteristics of the power supply when driving incandescent lampscapacitors, motors and etc.

In general, the overcurrent threshold current (I_(Thrs)) is defined as:I_(Thrs) =Q₁ ON V_(BE) /R₁₂ ≅0.6 volts/R₁₂. The threshold current shouldbe set approximately a factor of 2 higher than the power supply fullload current. The higher setting is to account for variations in thebase emitter voltage V_(BE) for the ON state of transistor Q₁ (Q₁ ONV_(BE)) resistor tolerances and to prevent false triggering of theovercurrent protection circuit 10.

The current through Q₁₁ will be a linear ramp as long as L₁₁ is linear.Note that the current through Q₁₁ will continue to ramp up even afterthe overcurrent comparator changes state signaling an overload current.Typically, Q₁₁ dows not turn OFF until a few microseconds after thecomparator U₁₁ changes state. This "coast-up" current will be smallerwith a faster switching transistor Q₁₁ and faster base drive circuit 14.

FIG. 2 shows a mechanization of the current limiter for two power switchtransistors (Q₁₁ and Q₁₂) connected in parallel. In this embodiment, theinvented circuit appears and operates the same as that of FIG. 1 withthe exception that the resistor R₁ has been replaced with two resistorsR₁ and R₃ each of which are roughly twice the size of resistor R₁ ofFIG. 1. This allows for the dual input into the transistor Q₁ whileretaining the RC time constant between resistors R₁ and R₃ andcompacitor C₁. For the same switching current, the parallel operationwill decrease the steady state on voltage drop by approximately a factorof 2.

In FIG. 2, resistors R₁₂ and R₂₀ serve two functions. The first is tosense the load current and the second is to help balance the steadystate load current between transistors Q₁₁ and Q₁₂. The transient lowcurrents cannot be balanced by the resistors R₁₂ and R₂₀ because one ofthe two transistors always switches faster. This means that during theinstant of switching ON, the fastest transistor carries 100% of theswitched ON current. Conversely, the slowest transistor carries 100% ofthe switched OFF current during the switching transient. If thetransistor switching time is made very short, the instant of 100%individual conduction (theoretically 100% overload) becomes so shortthat transistor junction (hot spot) damage is avoided provided there isadequate base drive.

A third embodiment is shown in FIG. 3. This embodiment is shown toillustrate that the invention can be realized through the use of asingle voltage regulation comparator U₁₂. Elimination of one comparatorsaves DC power and reduces the parts count for the overall circuit.Thus, the embodiment of FIG. 3 uses less stand-by power. This is animportant attribute for space applications where power conservationsimplicity and reliability are very important.

The previous discussion for the embodiments of FIGS. 1 and 2 applieshere for components R₁, R₂, C₁ and Q₁. In the embodiment of FIG. 3, thecapacitor C₁₀₁ in series with the capacitor C₁₀₂ corresponds with thecapacitor C₂ of the embodiment of FIGS. 1 and 2. The diode CR₁ providesthe `OR` function so that either the voltage sensing circuit 16 or theovercurrent limiter circuit 10 can control the same comparator U₁₂.

During normal operation (no overload) the diode CR₁ is back biased byabout 3 volts so that the overcurrent circuit does not load the voltagesensing divider consisting of resistors R₁₀₂ and R₁₀₃. These resistorsprovide a signal to the comparator U₁₂ which is compared to a referencepotential through resistors R₁₃ through R₁₇ as discussed above withregard to FIG. 1 to allow the comparator U₁₂ to provide signals to thebase drive circuit 14 for controlling the switching transistor Q₁₁.

When there is an overcurrent condition, the diode CR₁ overrides thesignal from the voltage sensing divider (R₁₀₂ and R₁₀₃) and transmitsthe linear current ramp from Q₁ to the capacitors C₁₀₁ and C₁₀₂. Thisoverride signal quickly changes the state of the comparator U₁₂ to turnOFF the switching transistor Q₁₁. When the transistor Q₁₁ goes OFFtransistor Q1 goes OFF.

The current through the diode CR₁ places a charge voltage approximatelyequal the hysteresis of the comparator on the series connectedcapacitors C₁₀₁ and C₁₀₂. In the event that the overvoltage andovercurrent thresholds occur at the same instant, the current throughthe diode CR₁ will not add much change voltage to the capacitors C₁₀₁and C₁₀₂. However, 1/2 cycle later (e.g., in about 25 microseconds) thevoltage sensing circuit 16 will turn the switching transistor Q₁₁ backON. Now diode CR₁ will act to quickly charge C₁₀₁ and C₁₀₂ to trip thethreshold of U₁₂ which turns Q₁ back OFF. That is, diode CR₁exponentially charges the capacitors C₁₀₁ and C₁₀₂ by the full amount ofthe comparator hysteresis which will turn OFF Q₁₁ within severalmicroseconds. Q₁₁ will now stay OFF until the delta voltage, quickcharged into the capacitors C₁₀₁ and C₁₀₂, decays. The decay path forthe capacitor C₁₀₁ is slow because it is via the parallel combination ofresistors R₁₀₂ and R₁₀₃. This quick charge, slow decay on C₁₀₁ and C₁₀₂forms a relaxation oscillator. The function of the relaxation oscillatoris the same as that discussed for FIG. 1. The frequency of therelaxation oscillator is set by the magnitude of the capacitors C₁₀₁ andC₁₀₂, the parallel resistance of the voltage divider provided byresistors R₁₀₂ and R₁₀₃, and the hysteresis voltage of U₁₂. The abovediscussion regarding the optimum frequency of this oscillator alsoapplies to the embodiment of FIG. 3. One kilohertz is typical.

The crossover network and the voltage sensing circuit 16 places somelimitations on the range of values for the capacitors 101 and C₁₀₂.However, a compromise can be reached so that both this overcurrentsensing circuit and the voltage sensing circuit 16 work well together.

                  TABLE I                                                         ______________________________________                                        Components          Specification                                             ______________________________________                                        Resistors           Ohms                                                      R.sub.1             100                                                       R.sub.2             100                                                       R.sub.3             220                                                       R.sub.4             10k                                                       R.sub.12            0.05                                                      R.sub.13            1.74k                                                     R.sub.14            115k                                                      R.sub.15            20.5k                                                     R.sub.16            28k                                                       R.sub.17            20.5k                                                     R.sub.18            17.4k                                                     R.sub.19            220                                                       R.sub.20            0.05                                                      R.sub.101           1k                                                        R.sub.102           64.9k                                                     R.sub.103           22.1k                                                     Capacitors          Farads                                                    C.sub.1             0.01μ                                                  C.sub.2             0.1μ                                                   C.sub.101           1μ                                                     C.sub.102           1μ                                                     C.sub.11            100 pico                                                  Inductors L.sub.11  Diodes                                                    CR.sub.1            1N4150                                                    CR.sub.11           1N3891                                                    Transistors                                                                   Q.sub.1             2N5680                                                    Q.sub.11            2N6287                                                    Q.sub.12            2N6287                                                    Amplifiers                                                                    U.sub.11            LM139                                                     U.sub.12            LM139                                                     ______________________________________                                    

While the present invention has been described herein with reference toparticular embodiments for a particular application, it is to beunderstood that the invention is not limited thereto. Those havingordinary skill in the art and access to the teachings of this inventionwill recognize modifications within the scope thereof. For example,while the above-described embodiments show hysteresis comparator typeswitching regulators, it is understood the invention is applicable toclocked comparator type switching regulators. It is contemplated by theappended claims to cover any and all such modifications.

What is claimed is:
 1. In a switching regulator power supply having a switching transistor for delivering power to a load connected to an output terminal thereof and control means for activating and deactivating said transistor on receipt of predetermined input signls; a circuit for controlling the switching transistor to limit the current therethrough whenever the load is substantially overloaded, said circuit comprising:first means for sensing the current through said switching transistor, including a current sensing resistor for developing a first voltage in response to the current through the switching transistor, said current sensing resistor being connected between a source of electrical energy and a terminal of the switching transistor and filter means for mitigating the effects of switching regulator transients, said filter means being connected to second means for providing to said control means a control signal whenever the current through said switching transistor exceeds a predetermined threshold, said second means including a current generator having a bipolar transistor biased for nominal operation in its active mode and being responsive to the current sensed in the switching transistor, means for developing a second voltage signal relative to ground level in response to the output of the current generator, and a threshold detector for comparing the second developed voltage to a reference potential and to generate said control signal to said control means to deactivate the switching transistor whenever the sensed current exceeds a predetermined threshold.
 2. The circuit of claim 1 wherein said means for developing said second voltage signal includes a capacitor connected between the current generator and ground.
 3. The circuit of claim 1 wherein said current generator includes means for providing parasitic oscillation suppression.
 4. In a switching regulator power supply having a switching transistor for delivering power to a load connected to an output terminal thereof and control means for activating and deactivating said transistor on receipt of predetermined input signals; a circuit for controlling the switching transistor to limit the current therethrough whenever the load is substantially overloaded, said circuit comprising:first means for sensing the current through said switching transistor including filter means for mitigating the effects of switching regulator transients, said filter means being connected to second means for providing to said control means a control signal whenever the current through said switching transistor exceeds a predetermined threshold, said second means including a current generator responsive to the current sensed in the switching transistor, means for developing a second voltage signal relative to ground level in response to the output of the current generator, and a threshold detector for comparing the second developed voltage to a reference potential and to generate said control signal for said control means to deactivate the switching transistor whenever the sensed current exceeds a predetermined level.
 5. The circuit of claim 4 wherein said first means includes a current sensing resistor connected between a source of electrical energy and a terminal of said switching transistor.
 6. The circuit of claim 4 wherein said current generator includes a bipolar transistor biased for nominal operation in its active mode.
 7. The circuit of claim 4 wherein said means for developing a voltage at ground level includes a capacitor connected between the current generator and ground.
 8. The circuit of claim 4 wherein said current generator includes means for providing parasitic oscillation suppression.
 9. In a switching regulator power supply having a switching transistor for delivering power to a load connected to an output terminal thereof and control means for activating and deactivating said transistor on receipt of predetermined input signals; a circuit for controlling the switching transistor to limit the current therethrough whenever the load is substantially overloaded, said circuit comprising:first means for sensing the current through said switching transistor including filter means for mitigating the effects of switching regulator transients, said filter means being connected to second means for providing to said control means a control signal whenever the current through said switching transistor exceeds a predetermined threshold.
 10. the circuit of claim 9 wherein said first means includes a current sensing resistor connected between a source of electrical energy and a terminal of said switching transistor. 